In conventional transmission of digital information the signal to be transmitted is protected against the effect of noise by the use of some sort of redundancy. The importance of coding as a way of improving the transmission reliability has been recognized from the very beginning of the digital communication era. At the receiving end the decision on a particular bit received is made on the basis of processing protected or coded signal, for example by considering repeated samples of the same piece of information. Assuming the noise affecting each sample is uncorrelated, then the total effect of the added redundancy reinforces the desired signal while averaging out the effect of the noise. The reliability of the decision made by the receiver is greatly increased when the signal to be transmitted is previously coded and this is clearly reflected as an increase in the performance of the digital link, e.g. as a reduced bit error rate. In general, channel coding provides the necessary protection against the degrading effect of Additive White Gaussian Noise (AWGN). This is carried out by exploiting temporal diversity, the principle used by conventional coding schemes.
When the effects of (fast) fading channels are taken into account, additional care is needed to keep performance figures comparable to those of a channel with only noise. Indeed, fading tends to modify adjacent transmitted bits approximately in the same proportion. It follows that any coding protection, including error correcting codes, will fail to protect the information transmitted when successive bits of a signal are affected by a highly correlated fading envelope. In order to exploit any conventional coding scheme the temporal structure of the bits to be transmitted has to be altered. In this fashion the correlated fading will now affect consecutive bits corresponding to non-successive signal bits. If the temporal structure is changed in a predetermined order, the received bits can be easily restructured by an inverse operation to produce a bit stream in the same order as the originally transmitted. Then channel decoding can take place as usually. The above time domain operations are known as (bit) interleaving at the transmitting end and deinterleaving at the receiving end. Interleaving is usually carried out by writing the coded bits into a matrix in a row-wise fashion and reading the bits to be transmitted in a column-wise fashion. It should be noted that the interleaving-deinterleaving operation imposes an inherent delay to the signal. For an r×q interleaving matrix, the signal will be delayed by an equivalent time corresponding to rq bits. In slowly changing environments, the fading envelope will be correspondingly slow and conversely, the coherence time will be large. The slower the fast-fading envelope, the more consecutive bits are involved (or correlated) with the fading and the larger should be the interleaving depth to provide effective protection.
In radio environments characterized by low mobility, e.g. indoor cells, the coherence time of the channel is large, typically at least hundreds of milliseconds. Coherence time reflects the change speed of the channel, and it can be said that the coherence time defines how far apart the bits should be placed in interleaving, so that their cross correlation is small enough in case of a fading channel. The fading depths are correspondingly long and thus the large interleaving depth required to provide enough protection could result in excessively long delays. Many applications are delay sensitive in the sense that large delays will affect the quality of the transmission. This is directly related with real-time applications like speech and video traffic, where long delays could be unacceptable. In addition, system constraints may also limit the maximum amount of used interleaving depth, especially with high-bit-rate users. In fact, long delays in conjunction with high-bit-rate traffic could require excessively large signal buffering capabilities at both ends. In summary, in low mobility radio environments the use of interleaving implies unacceptably long processing delays in many cases. The problem becomes more serious for higher bit rates. The importance of this problem can be seen from the fact that the above scenarios are typical for WCDMA (Wide-band Code Division Multiple Access) systems operating in pico/micro cells.
The envelope delay correlation of the signal received by the mobile station can be approximated by:ρ(τ)=J20(βvτ),  (1)
where τ is the time delay of interest, J0 is the Bessel function, v is the mobile station velocity and β=2π/λ, λ being the wavelength of the signal. The first zero (corresponding to the minimum delay with zero correlation) of the Bessel function occurs for an argument of approximately 2.4. As an example, the delay correlation (this figure is comparable to the coherence time of the channel) for a 2 GHz radio channel where a mobile station is nearly stationary or moves very slowly (v<1 Km/h) is larger than about 200 milliseconds. Assuming that an interleaving depth of the same order is used, then for a 2M bit/s signal at least some 400K signal bits have to be buffered. This could easily be well beyond the processing capacity of the terminal equipment.
The interleaving is utilized primarily to decorrelate the effect of fast fading on contiguous bits of information. However, an equivalent effect can be achieved by transmitting these adjacent bits from different antennas. The decorrelating effect achieved with the temporal interleaving can be replaced or extended in principle by using equivalent spatial processing or spatial interleaving. It should be noted that in order to obtain a duality in the spatial and temporal behaviour, the transmitting antennas have to produce correspondingly uncorrelated signals at the receiving end. Fortunately, this is quite true in pico/micro cells due to their typically large angular spreads, also in macrocells if antenna separation is adequately large or polarization diversity is applied. The same principle expressed by equation 1 can be extended to the spatial domain, in order to obtain the following space correlation expression:                                           ρ            ⁢                                                  ⁢                          (              d              )                                =                                    J              0              2                        ⁢                                                  ⁢                          (                              β                ⁢                                                                  ⁢                d                            )                                      ,                            (        2        )            
where d represents spatial separation. Again, the separation for zero correlation results in a required spatial separation d=0.38λ (βd=2.4), assuming the classical Doppler spectrum. It is important to emphasize that equivalent decorrelating effects can be obtained in the time domain when ρ(τ)=0 (equation 1) and in the space domain when ρ(d)=0 (equation 2).